K-BAND QUADRATURE MIXERS WITH PLASTIC-PACKAGED DIODES Matjaz Vidmar Quadrature (IQ) mixers are important signal-processing components in radio transmitters (as vector modulators), in radio receivers (as image-reject mixers and/or demodulators) as well as in other microwave equipment (phase-locked loops). A quadrature (IQ) mixer requires two balanced mixers that are precisely matched in amplitude and are operated at an accurate 90-degree phase shift. Precise amplitude matching and exact phase shifts are increasingly more difficult to obtain at higher frequencies. At K-band frequencies, the best technical solution is probably monolithic integration as presented in [1] and [2]. However, there are other possible hardware implementations that should not be neglected. In this article two practical K-band, quadrature-mixer designs are presented, using inexpensive, packaged semiconductor devices installed on conventional printed-circuit boards. Although most packaged semiconductor devices were developed for operation at relatively low frequencies, usually below 3GHz for the mass mobile-phone market, the small package size and excellent chip characteristics allow useful performance even in the K-band frequency range. The first application is a 20Gbps clock recovery that requires a quadrature phase detector in its phase-locked loop. The second application is a direct-conversion (zero-IF) 24GHz (ISM band) radio transceiver, that requires quadrature mixers both in the transmitter modulator as well as in the receiver demodulator. Finally a practical test method for quadrature mixers is presented that allows quick and efficient circuit optimization as well as verification of all important quadrature-mixer parameters. Antiparallel diode mixer ------------------------ At K-band and higher frequencies, the antiparallel diode mixer is a popular solution, since it requires a subharmonic local oscillator at one half of the conversion frequency. In other words, the antiparallel diode mixer has a built-in, local-oscillator frequency doubler. On the other hand, the antiparallel mixer insertion loss and noise figure are worse than those of fundamental-LO mixers, the IMD performance is also worse and the antiparallel mixer is much more sensitive to the LO signal level than fundamental-LO mixers. The principle of operation of an antiparallel mixer is shown on Fig. 1. The I(V) curve of a perfect antiparallel diode pair is an odd function that can be represented with a polynomial with just odd powers of V (C1, C3, C5, C7 etc). One of the third-order conversion products, RF input minus twice LO, is in fact the desired conversion product of a mixer with a subharmonic LO. Of course, other third-order and higher-order conversion products cause intermodulation distortion (IMD). In an antiparallel mixer, the highest conversion efficiency is obtained when each diode is conducting for about 90 electrical degrees of the LO signal. If the LO level is too low, the conversion efficiency may drop to zero if the diodes are not turned on at all. On the other hand, if the LO signal level is too high, either one or the other diode are turned on most of the time, again resulting in little impedance change and thus poor conversion efficiency. Conventional silicon schottky diodes typically require a LO level between +0dBm and +10dBm for best conversion efficiency in an antiparallel mixer in a 50-ohm environment. A frequently forgotten, but important attribute of the antiparallel mixer is that it is a balanced mixer. Thanks to the odd transfer function of a perfect antiparallel diode pair, rectification and second-harmonic generation are not possible. Even more important at very high frequencies, the mixer balancing of an antiparallel mixer is obtained without critical and complex baluns. Good mixer balancing is important in a phase detector to reduce the output offset voltage. Good mixer balancing is also important in a receiving mixer to suppress LO noise, especially at very low intermediate frequencies or in direct-conversion receivers. Finally, good mixer balancing is also required in a vector modulator in a transmitter to suppress the residual carrier. Since the RF, LO and IF frequencies are widely apart and no baluns are required, antiparallel mixer circuits are usually simple as shown on Fig. 2. At low IF values, the RF frequency is close to the second harmonic of the LO frequency. The conventional antiparallel mixer circuit (A) therefore includes just an open stub and a shorted stub. Both stubs are one quarter wavelength at the LO frequency and one half wavelength at the RF frequency. The IF is coupled through a simple LC lowpass. Unless two monolithically-integrated diodes are used as the antiparallel pair, package and circuit parasitics may corrupt the mixer balancing. When using single-packaged diodes at K-band frequencies, a completely symmetrical circuit is required for good mixer balancing as shown on Fig. 2 (B). An additional advantage of the latter is that no critical connections to ground (vias in the microstrip board) are required. The IF can be taken either from the LO side as shown on Fig. 2 (B) or from the RF side of the mixer. In both cases an IF return to ground has to be provided on the other side of the mixer. The quarter-wavelength RF traps introduce a substantial mismatch for the LO signal, so additional LO matching stubs are always required. Little if any matching is required on the RF side of the mixer, since the quarter-wavelength LO traps behave as half-wavelength resonators at the input RF frequency. Single silicon schottky diodes BAT14-02W, manufactured by Infineon Technologies (formerly Siemens Semiconductors) are used in the described K-band antiparallel mixer. The BAT14-02W is a medium-barrier diode (430mV forward drop at 1mA) and is packaged in the very small SCD80 plastic SMD package. An antiparallel mixer with two diodes BAT14-02W requires about +7dBm of LO power for operation at K-band frequencies. 20Gbps RZ clock recovery ------------------------ In optical-fiber communications, specially shaped lightwave pulses called solitons are frequently used for data rates beyond 10Gbps. In the receiving photodiode the soliton pulses are converted in a Return-to-Zero (RZ) electrical data stream. RZ-encoded data has a very strong, discrete spectral line at the clock frequency. A RZ clock recovery can be simply a narrow bandpass filter, tuned to the clock frequency. A phase-locked loop (PLL) is frequently used as a narrowband filter in clock-recovery circuits. A suitable second-order feedback network in the PLL can remove any static clock-phase errors, regardless of component tolerances and clock-frequency offset. Unfortunately, the initial frequency offset of such a PLL may be much larger than the acquisition bandwidth of the PLL. A search logic together with a reliable lock detector are therefore required in a PLL-based clock recovery. A clock recovery PLL requires two phase detectors operating in quadrature. The in-phase (I) detector provides the lock information while the quadrature (Q) detector provides the loop feedback. In the unlocked state, the search logic scans the whole frequency range of the VCO. As soon as lock is detected, the feedback loop is closed by connecting the Q detector output to the VCO control input. The circuit diagram of a practical 20GHz quadrature phase detector is shown on Fig. 3. The circuit includes a two-stage, selective RF preamplifier and two antiparallel mixers with BAT14-02W diodes. Both mixers are fed in phase with the 10GHz LO signal. A 90-degree phase shift is introduced by the RF-input line to the Q mixer being one-quarter wavelength longer than the corresponding line to the I mixer. Both RF and LO signals are split with Wilkinson dividers to reduce the crosstalk between the two mixers. The selective RF preamplifier has a gain of about 13dB including the 20GHz bandpass insertion loss. The latter is required to prevent overloading both mixers with the wideband signal coming from the photodiode. A RF input of -10dBm at 20GHz produces I and Q outputs of about +/-150mV on a high-impedance load with a nominal +10dBm LO drive at 10GHz. The 20GHz quadrature phase detector is built as a microstrip circuit, etched on 0.5mm (19mils) thick Ultralam 2000 laminate (Rogers Corporation) with a dielectric constant of 2.43. The circuit pattern is shown on Fig. 4 and has the dimensions of 80mm X 30mm. Since the circuit is a prototype, there are no plated-through holes. All components are grounded through 3.2mm (1/8") diameter holes in the laminate that are covered with 0.1mm-thick copper foil on the groundplane side. The 1pF capacitors are made each with two 0.5pF, 0402-size ceramic capacitors installed in parallel to minimize the discontinuities on the microstrip lines. All resistors are 0805 size. The resistors in the RF part of the circuit (the two 330ohm resistors in the gates of the FETs and the two 100ohm resistors in the Wilkinson dividers) are installed "upside-down" with the resistive layer facing the surface of the printed-circuit board. The block diagram of the 20Gbps RZ (soliton) clock recovery is shown on Fig. 5. The clock recovery includes a high-speed photodiode, an additional 20GHz narrowband amplifier, a 20GHz quadrature phase detector, a PLL search logic, a PLL loop amplifier and a 2.5GHz VCO followed by phase shifters and frequency multipliers. The EPM820FJ-S high-speed photodiode (EPITAXX) includes an internal 50ohm termination and is specified for 15GHz bandwidth. The expected optical input sensitivity should be around -5dBm. Depending on the shape of the input (soliton) optical pulses, even better sensitivities were observed in practice. Due to the large impedance mismatch between the high-impedance photodiode and the 50-ohm system, a -5dBm optical input only produces a RF signal of about -35dBm at 20GHz, so an additional amplifier is required in front of the phase detector. The remaining circuits of the 20Gbps clock recovery are designed around the same phase shifters and other circuits described in [3]. The VCO is operated at 2.5GHz to allow simplier phase shifters followed by frequency multipliers. The 20Gbps clock recovery supplies two main 10GHz clock outputs for the 20/10Gbps demultiplexer (D-flip-flop) and for the 10Gbps error detector. Both 10GHz clock outputs have independent phase shifters. An additional 2.5GHz output is provided as a trigger for a sampling oscilloscope. Finally, an additional phase shifter is built in the PLL itself, supplying the 10GHz local oscillator to the 20GHz quadrature phase detector. This additional phase shifter is very convenient when the optical-fiber length in front of the clock recovery is changed while all other optical and microwave connections remain unchanged. The described clock-recovery design was intended for particular laboratory experiments that asked for different output options. Although a direct 20GHz output could easily be implemented, it was not provided since it was not required for the abovementioned experiments. 24GHz direct-conversion transceiver ----------------------------------- Direct-conversion transceivers include a direct vector modulator in the transmitter and downconversion to a very low intermediate frequency in the same range as the baseband modulation signal in the receiver. Critical and expensive microwave hardware like narrow bandpass filters is not required since there are no image responses or other close-in spurious signals to be filtered out. A direct-conversion or zero-IF transceiver is therefore a promising solution for no-tune, low-cost mass-market applications if simple and reliable quadrature (IQ) mixers are available for the frequency range of interest. In direct-conversion or zero-IF transceivers, the subharmonic-LO mixer has yet another advantage. Since the subharmonic local oscillator operates on a different frequency from the RF front-end of the transceiver, much less shielding is required to prevent LO leakage from being picked up by the receiving antenna. Less shielding is also required to prevent the transmitted signal from disturbing the LO source. Poor LO shielding, resulting in an interaction with the RF front-end of the transceiver, corrupts the balance and quadrature of both receiving and transmitting mixers. The latter is usually a major drawback of direct-conversion designs. To check the feasibility of direct-conversion designs, a Weaver SSB analog voice transceiver for 24GHz was designed, built and tested. The transceiver includes quadrature mixers with antiparallel diodes both in the transmitter and in the receiver. Of course, a similar transceiver design could also be used for BPSK or QPSK data transmission in the 24GHz ISM band. The design of the 24GHz quadrature receiving mixer, shown on Fig. 6, is very similar to the already described 20GHz phase detector. Two simple 12GHz bandpass filters are added to prevent low-frequency noise from the local oscillator from disturbing the weak I and Q IF signals. Two IF preamplifiers with two BF199 transistors (Philips) are included in the quadrature-mixer module for a similar reason. Both IF preamplifiers receive their supply voltage from the following quadrature IF amplifier. The 24GHz RF preamplifier is also similar to the 20GHz version. Unfortunately the ATF35076 PHEMTs are only specified up to 18GHz by their manufacturer [4]. While it is relatively easy to match these devices at 20GHz, it is much more difficult to obtain some useful gain from them at 24GHz, probably due to an internal package resonance. Tuning stubs at both input and output are required to obtain a similar gain as in the 20GHz version. Both transistors operate at a zero gate bias for the maximum gain. The gate bias network includes a lowpass and a 68ohm resistor to decrease the gain at lower frequencies. Red LEDs are used as very-low-noise shunt regulators for the drain supplies of both transistors. Supply regulators are required since the 24GHz transceiver supply is +12V. The 24GHz quadrature receiving mixer is built as a microstrip circuit, etched on 0.5mm (19mils) thick Ultralam 2000 laminate (Rogers Corporation) with a dielectric constant of 2.43. The circuit pattern is shown on Fig. 7 and has the dimensions of 80mm X 30mm just like the 20GHz phase detector. The only difference in the construction from the 20GHz phase detector are two 1nF feedthrough capacitors installed in 3.2mm (1/8") diameter holes in the substrate. The tuning stubs are small pieces of 0.1mm-thick, pretinned copper foil. The noise figure of the described 24GHz quadrature mixer is rather high. Experiments have shown that most noise comes from the 12GHz local oscillator, especially with a very low IF as in the case of a Weaver SSB voice receiver. In addition to the built-in, two-stage preamplifier a LNA gain of at least 25dB is required to mask the local-oscillator noise. The 24GHz quadrature transmitting mixer (vector modulator) is identical to the receiving mixer. The 12GHz LO signal level is raised to about +14dBm to obtain a higher output power at 24GHz. At higher LO levels, the BAT14-02W diodes generate high-order harmonics up in the millimeter-wave region. High-order harmonics corrupt the mixer balancing and may even cause interaction between the two antiparallel mixers. In the latter case, a very simple countermeasure may be to swap the two mixer diodes of one antiparallel mixer to flip the phase of the offending harmonic. The transmitting mixer includes a two-stage selective RF amplifier at 24GHz. The latter is identical to the RF preamplifier in the receiving mixer except that the input and output are swapped. An undistorted SSB signal of about -3dBm at 24GHz is obtained at the output of the two-stage RF amplifier. Of course, the whole transceiver includes additional amplifier stages to raise the transmitter output power as well as improve the receiver noise figure. Testing quadrature mixers ------------------------- The specification of a quadrature (IQ) mixer requires several parameters to be tested. Besides the parameters of the single mixers, precise 90-degree phase offset and gain tracking between the two mixers are also required. A simple and quick test procedure to display all important quadrature-mixer parameters is also required for tuning or optimizing a circuit during development. The mixer test setup suggested on Fig. 8 can be used to characterize phase detectors or receiving mixers with a very low IF. Very low intermediate frequencies, comparable to the baseband modulation signals, are used in direct-conversion transceivers. The same setup can also provide useful data for transmitting mixers, since most interesting mixer parameters are reciprocal under small RF/IF signal conditions. While working with mixers, a suitable LO source is usually available as part of the equipment where the mixer is going to be used. A CW signal generator equipped with a variable attenuator is also a standard piece of test equipment for testing receivers. The only additional item on Fig. 8 is a low-frequency XY oscilloscope display. The bandwidth of the latter should include the expected intermediate-frequency range, usually less than 1MHz in direct-conversion transceivers. The RF signal generator is tuned to a frequency that has a small offset from the appropriate local-oscillator harmonic, depending on the type of mixer. The XY oscilloscope display then shows a rotating bright spot. The direction of rotation, clockwise or counterclockwise, depends on whether the RF signal generator is tuned below or above the appropriate LO harmonic. While making measurements on K-band mixers, a frequency offset between 100Hz and 10kHz is usually a practical choice. If the frequency offset is too low, the phase noise of both RF and LO signal sources makes the oscilloscope trace very noisy. If the frequency offset is too high, the frequency response of the IF network and XY display may corrupt the result. In all cases it is very useful to check the XY display across the expected frequency range before making any real measurements. A perfect quadrature (IQ) mixer should display a perfect circle centered on the XY screen. Real quadrature mixers display an ellipse due to gain differences between the I and Q channels and phase offset differing from the ideal 90 degrees. Finally, the center of the ellipse is offset from the center of the XY display due to poor mixer balance as shown on Fig. 9. Antiparallel mixers with medium-barrier schottky diodes like the BAT14-02W typically provide a linear output range of about +/-150mV on a high-impedance load. If the RF signal level is further increased, mixer saturation occurs. The ellipse on the XY display becomes distorted. The first visible effect is usually an egg-shaped curve due to slight limiting in the mixer. Poor quadrature-mixer balance can be described with the ellipse-center offset phasor "o". Poor antiparallel-mixer balance is caused by single-diode tolerances as well as high-order harmonics at high LO drive levels. At a LO drive level of +10dBm for the whole quadrature mixer (+7dBm per antiparallel mixer), the unbalance phasor magnitude was in the 15mV range. In the transmit mixer, a residual carrier suppression of about -20dB was measured at +14dBm LO drive in a different test setup, confirming that reciprocity works in this case. The mixer balance could be improved by selecting matched pairs of the BAT14-02W mixer diodes. It should be noted that the BAT14-02W diodes used in the experiments were the very first pre-production samples obtained from Infineon. Fine tuning of the mixer balance can be adjusted by precise positioning of the diodes on the printed-circuit board. Since the BAT14-02W diodes are soldered to the printed-circuit board as conventional SMD components, they can be repositioned by simple soldering operations. The most important parameter of a quadrature (IQ) mixer is probably the axial ratio "R" of the ellipse observed on the XY oscilloscope display. The axial ratio defines the image (or unwanted-sideband) rejection "A" of a quadrature mixer. If the inclination angle "Alpha" of the major axis of the ellipse is also known, the exact cause of the ellipticity can be found: gain error, phase error or both. The above information is very useful while developing or tuning the quadrature-mixer circuit. A major-axis angle close to 45 degrees or 135 degrees indicates that the phase offset between the I and Q mixers differs from the ideal 90 degrees. A major-axis angle close to 0 degrees or 90 degrees indicates a gain difference between the I and Q channels. Most important of all, the effects of additional tuning stubs can be observed immediately on the XY oscilloscope display. The designs of both K-band quadrature mixers presented in this article were optimized with the above test procedure. The final designs achieved an axial ratio of about 1dB, corresponding to an image (or unwanted-sideband) rejection of about -25dB. Due to the relatively thick microstrip board (0.5mm teflon at K-band) the unwanted effects due to microstrip radiation and reflections from the walls of the shielding enclosure were in the same order of magnitude. Conclusion ---------- Two practical K-band quadrature (IQ) mixers are presented as part of a 20Gbps clock recovery or as components of a 24GHz direct-conversion radio transceiver. Both designs use inexpensive, plastic-packaged SMD diodes soft-soldered on inexpensive softboard substrates. The single mixers use antiparallel diodes to acheive subharmonic LO operation, good mixer balancing and good residual carrier suppression. A simple and efficient method for testing quadrature IQ mixers is also presented. The described K-band IQ mixers achieve an image (or unwanted-sideband) suppression of about -25dB without additional tuning. References: ----------- [1] Ali E. Ashtiani, Tacar Gokdemir, Georgios Passiopoulos, Ali Rezazadeh, Suengil Nam, Ian Robertson, "Miniaturized Low Cost 30GHz Monolithic Balanced BPSK and Vector Modulators: Part I", Microwave Journal, vol. 42, no. 3, March 1999, pp 100-108. [2] Ali E. Ashtiani, Tacar Gokdemir, Georgios Passiopoulos, Ali Rezazadeh, Suengil Nam, Ian Robertson, "Miniaturized Low Cost 30GHz Monolithic Balanced BPSK and Vector Modulators: Part II", Microwave Journal, vol. 42, no. 4, April 1999, pp 104-114. [3] Matjaz^H~ Vidmar, "Microstrip Resonant Phase Shifters", Microwave Journal, vol. 42, no.9, September 1999, pp 127-136. [4] Hewlett Packard, "Communications Components Designer's Catalog, GaAs and Silicon Products", 1993, pp 7.73-7.75. *************************************************************** List of figures: ---------------- Fig. 1 - Antiparallel diode pair perfomance. Fig. 2 - Antiparallel mixer circuits. Fig. 3 - 20GHz quadrature phase detector. Fig. 4 - 20GHz phase detector microstrip board. Fig. 5 - 20Gbps RZ (soliton) clock recovery. Fig. 6 - 24GHz quadrature receiving mixer. Fig. 7 - 24GHz receiving mixer microstrip board. Fig. 8 - Testing quadrature mixers with subharmonic LO. Fig. 9 - Quadrature mixer performances from ellipse parameters.